High power battery powered RF amplifier topology

ABSTRACT

In various embodiments, a control circuit for a radio frequency drive of an electrosurgical device is disclosed comprising a voltage data input configured to receive voltage data, a current data input configured to receive current data, and a switching signal output configured to source a switching signal. The control circuit is configured to adjust a frequency of the switching signal based on the voltage data and the current data. The radio frequency drive comprising a transformer comprising a first tap, a second tap, a third tap, a first portion of a primary coil, a second portion of the primary coil, and a secondary coil. Two of the first, second, and third taps are selected to drive the primary coil between the two selected taps.

CROSS-REFERENCE TO RELATED APPLICATION

The present application is a continuation application claiming priorityunder 35 U.S.C. § 120 to U.S. patent application Ser. No. 14/579,543,entitled HIGH POWER BATTERY POWERED RF AMPLIFIER TOPOLOGY, filed Dec.22, 2014, which issued on Dec. 25, 2018 as U.S. Pat. No. 10,159,524, theentire disclosure of which is hereby incorporated by reference herein.

INTRODUCTION

The present disclosure relates to the design of high-powerradiofrequency amplifiers for use in electrosurgical instruments thatemploy radiofrequency energy to cauterize or coagulate tissue.

Conventional corded electrosurgical instruments are large in size, havelarge power supplies and control electronics, and take up a lot of spacein the operating room. Corded electrosurgical instruments areparticularly cumbersome and difficult to use during a surgical procedurein part due to tethering of the hand-held electrosurgical instrument tothe power supply and control electronics and the potential for cordentanglement. Some of these deficiencies have been overcome by providingbattery powered hand-held electrosurgical instruments in which the powerand control electronics are mounted within the instrument itself, suchas within the handle of the instrument, to reduce the size of theelectrosurgical instrument and make such instruments easier to useduring surgical procedures.

Electrosurgical medical instruments generally include an end effectorhaving an electrical contact, a radiofrequency (RF) generation circuitfor generating an RF drive signal and to provide the RF drive signal tothe at least one electrical contact where the RF generation circuit alsoincludes a resonant circuit. The RF circuit includes circuitry togenerate a cyclically varying signal, such as a square wave signal, froma direct current (DC) energy source and the resonant circuit isconfigured to receive the cyclically varying signal from the switchingcircuitry. The DC energy source is generally provided by one or morebatteries that can be mounted in a handle portion of the housing of theinstrument, for example.

Batteries mounted within the electrosurgical instrument have severallimitations. For example, the amount of power the batteries provide mustbe balanced against their weight and size. Thus electrosurgicalinstruments employing RF energy typically include a high-power RFamplifier, for instance, one producing 5A RMS output, 300 W, at 170VRMS.

The design of high-power RF amplifiers, specifically the design of theoutput transformer, is complicated by the relatively low voltageprovided by battery packs that are practical for handheld devices meantfor possibly delicate uses. Such battery packs usually provide voltagein multiples of 4.2V (i.e. LiIon cell potential). The upper practicallimit for handled devices is up to five cells in series—for example, ina 2P5S (sets of two parallel cells, five sets in a series string)—due tospace and weight constraints. Even with a 2P5S battery configuration,selecting a workable turns ratio for the output transformer is at best acompromise between the maximum allowable primary current when in currentlimit mode and the need for a reasonable turns ratio for generating170-250V RMS in the voltage control region of an electrosurgicaldevice's power curve. The present disclosure provides a compact,optimally performing high power RF amplifier with significantly lesscompromise in the design of the output transformer. The presentdisclosure provides systems and methods for changing the turns ratio atwill, synchronously with the carrier frequency of the energy device.Thus it is possible to adapt to the requirements of each region of anelectrosurgical or ultrasonic power curve (current limit, power limitand voltage limit).

SUMMARY

In various embodiments, a control circuit for a radio frequency drive ofan electrosurgical device is disclosed. The control circuit comprises avoltage data input configured to receive voltge data from a voltagesensing circuit, a current data input configured to receive current datafrom a current sensing circuit, and a switching signal output configuredto source a switching signal to the radio frequency drive of theelectrosurgical device. The control circuit is configured to adjust afrequency drive of the electrosurgical device. The control circuit isconfigured to adjust a frequency of the switching signal based on thevoltage data and the current data. The radio frequency drive comprises atransformer. The transformer comprises a first tap including a firsthalf bridge driver, a second tap including a second half bridge driver,a third tap including a third half bridge driver, a first portion of aprimary coil located between the first tap and the second tap, a secondportion of the primary coil located between the second tap and the thirdtap and the second tap, a second portion of the primary coil locatedbetween the second tap and the third tap, and a secondary coil. Thefirst, second, and third half bridge drivers are configured toselectively turn on or turn off the first, second, and third taps,respectively. Two of the first, second, and third taps are selected todrive the primary coil between the two selected taps, which allows thetransformer to provide a plurality of winding ratio values. A number ofcoil turns of the primary coil between the two selected taps and anumber of coil turns of the secondary coil determine an overall windingratio value of the transformer. The overall winding ratio value is oneof the plurality of winding ratio values provided by the transformer.

In various embodiments, a method of controlling a radio frequency drivecircuit of an electrosurgical device is disclosed. The method comprisessampling, by a synchronous I/Q sampling circuit, a plurality of outputvoltage values of an output voltage of the electrosurgical device,sampling, by the synchronous I/Q sampling circuit, a plurality of outputcurrent values of an output current of the electrosurgical device, andsourcing, by a square wave generation module, a switching signal todrive the radio frequency drive circuit. The output voltage and theoutput current are sourced by the radio frequency drive circuitcomprising a transformer. The transformer further comprises a first tapincluding a first half bridge driver, a second tap including a secondhalf bridge driver, a third tap including a third half bridge driver, afirst portion of a primary coil located between the first tap and thesecond tap, a second portion of the primary coil located between thesecond tap and the third tap, and a secondary coil, wherein the first,second, and third half bridge drivers are configured to selectively turnon or turn off the first, second, and thrid taps, respectively. Two ofthe first, second, and third taps are selected to drive the primary coilbetween the two selected taps, which allows the transformer to provide aplurality of winding ratio values. A number of coil turns of the primarycoil between the two selected taps and a number of coil turns of thesecondary coil determine an overall winding ratio value of thetransformer. The overall winding ratio value is one of the plurality ofwinding ratio values provided by the transformer.

In another embodiment, the upper and lower switch elements comprisesolid-state switching elements. In another embodiment, the solid-stateswitching elements comprise MOSFETs.

In another embodiment, the upper and lower switch elements compriseIGBTs.

In another embodiment, the upper and lower switch elements comprisemechanical relays.

In another embodiment the current amplifier comprises a parallelcapacitor on the secondary coil, such that the output produced by theamplifier is a sine wave.

In another embodiment, the minimum winding ratio is 4:1.

In another embodiment, the maximum winding ratio is 15:1.

In one embodiment, an electrosurgical medical instrument comprises aradio frequency (RF) generation circuit coupled to and operated by abattery and operable to generate an RF drive signal and to provide theRF drive signal to at least one electrical contact, wherein the RFgeneration circuit comprises: A current amplifier, comprising atransformer, the transformer comprising one or more taps on the primarycoil, wherein each tap comprises a half bridge driver.

In another embodiment, the half bridge driver comprises an upper switchelement and a lower switch element, a high-side drive input connected tothe input of the upper switch element, a low-side drive input connectedto the input of the lower switch element, wherein the upper and lowerswitch elements are connected in a cascade arrangement and the output ofthe half bridge driver is taken from the node between the upper andlower switch elements.

In another embodiment, the upper and lower switch elements comprisesolid-state switching elements. In another embodiment, the solid-stateswitching elements comprise MOSFETs.

In another embodiment, the upper and lower switch elements compriseIGBTs.

In another embodiment, the upper and lower switch elements comprisemechanical relays.

In another embodiment, the medical instrument comprises a parallelcapacitor on the secondary coil, such that the output produced by theamplifier is a sine wave.

In another embodiment, the minimum winding ratio is 4:1.

In another embodiment, the maximum winding ratio is 15:1.

In one embodiment, a current amplifier, comprises a transformercomprising one or more taps on the primary coil, wherein each tapcomprises a half bridge driver configured to selectively turn on or turnoff the tap, wherein the half bridge driver comprises an upper switchelement and a lower switch element, a high-side drive input connected tothe input of the upper switch element, a low-side drive input connectedto the input of the lower switch element, wherein the upper and lowerswitch elements are connected in a cascade arrangement and the output ofthe half bridge driver is taken from the node between the upper andlower switch elements; and a parallel capacitor on the secondary coil,such that the output produced by the amplifier is a sine wave; whereinthe minimum winding ratio is 4:1; and wherein the maximum winding ratiois 15:1.

In another embodiment, the upper and lower switch elements comprisesolid-state switching elements.

FIGURES

The novel features of the embodiments described herein are set forthwith particularity in the appended claims. The embodiments, however,both as to organization and methods of operation may be betterunderstood by reference to the following description, taken inconjunction with the accompanying drawings as follows:

FIG. 1 illustrates one embodiment of an RF amplifier with one or more oftaps on the primary coil, wherein each tap is controlled by a halfbridge driver;

FIG. 2 illustrates one embodiment of a half bridge circuit that may beemployed by the half bridge driver;

FIG. 3 illustrates an RF drive and control circuit, according to oneembodiment;

FIG. 4 illustrates a perspective view of one embodiment of thetransformer shown as transformer in connection with the RF drive circuitillustrated in FIG. 3;

FIG. 5 illustrates a perspective view of one embodiment of the primarycoil of the transformer illustrated in FIG. 4;

FIG. 6 illustrates a perspective view of one embodiment of a secondarycoil of the transformer illustrated in FIG. 4;

FIG. 7 illustrates a bottom view of the primary coil illustrated in FIG.5;

FIG. 8 illustrates a side view of the primary coil illustrated in FIG.5;

FIG. 9 illustrates a sectional view of the primary coil illustrated inFIG. 5 taken along section 28-28;

FIG. 10 illustrates a bottom view of the secondary coil illustrated inFIG. 6;

FIG. 11 illustrates a side view of the secondary coil illustrated inFIG. 6;

FIG. 12 illustrates a sectional view of the secondary coil illustratedin FIG. 11 taken along section 31-31;

FIG. 13 is a perspective view of one embodiment of the inductor shown asinductor L_(s) in connection with the RF drive circuit illustrated inFIG. 3;

FIG. 14 illustrates a bottom view of the inductor illustrated in FIG.13;

FIG. 15 illustrates a side view of the inductor illustrated in FIG. 13;

FIG. 16 illustrates a sectional view of the inductor illustrated in FIG.15 taken along section 35-35.

FIG. 17 illustrates the main components of the controller, according toone embodiment;

FIG. 18 is a signal plot illustrating the switching signals applied tothe FETs, a sinusoidal signal representing the measured current orvoltage applied to the load, and the timings when the synchronoussampling circuitry samples the sensed load voltage and load current,according to one embodiment; and

FIG. 19 illustrates a drive waveform for driving the FET gate drivecircuitry, according to one embodiment.

DESCRIPTION

In the following detailed description, reference is made to theaccompanying drawings, which form a part hereof. In the drawings,similar symbols and reference characters typically identify similarcomponents throughout the several views, unless context dictatesotherwise. The illustrative embodiments described in the detaileddescription, drawings, and claims are not meant to be limiting. Otherembodiments may be utilized, and other changes may be made, withoutdeparting from the scope of the subject matter presented here.

The following description of certain examples of the technology shouldnot be used to limit its scope. Other examples, features, aspects,embodiments, and advantages of the technology will become apparent tothose skilled in the art from the following description, which is by wayof illustration, one of the best modes contemplated for carrying out thetechnology. As will be realized, the technology described herein iscapable of other different and obvious aspects, all without departingfrom the technology. Accordingly, the drawings and descriptions shouldbe regarded as illustrative in nature and not restrictive.

It is further understood that any one or more of the teachings,expressions, embodiments, examples, etc. described herein may becombined with any one or more of the other teachings, expressions,embodiments, examples, etc. that are described herein. Thefollowing-described teachings, expressions, embodiments, examples, etc.should therefore not be viewed in isolation relative to each other.Various suitable ways in which the teachings herein may be combined willbe readily apparent to those of ordinary skill in the art in view of theteachings herein. Such modifications and variations are intended to beincluded within the scope of the claims.

Before explaining the various embodiments of the high power batterypowered RF amplifier technology in detail, it should be noted that thevarious embodiments disclosed herein are not limited in theirapplication or use to the details of construction and arrangement ofparts illustrated in the accompanying drawings and description. Rather,the disclosed embodiments may be positioned or incorporated in otherembodiments, variations and modifications thereof, and may be practicedor carried out in various ways. Accordingly, embodiments of the surgicaldevices disclosed herein are illustrative in nature and are not meant tolimit the scope or application thereof. Furthermore, unless otherwiseindicated, the terms and expressions employed herein have been chosenfor the purpose of describing the embodiments for the convenience of thereader and are not to limit the scope thereof. In addition, it should beunderstood that any one or more of the disclosed embodiments,expressions of embodiments, and/or examples thereof, can be combinedwith any one or more of the other disclosed embodiments, expressions ofembodiments, and/or examples thereof, without limitation.

For clarity of disclosure, the terms “proximal” and “distal” are definedherein relative to a human or robotic operator of the surgicalinstrument. The term “proximal” refers the position of an element closerto the human or robotic operator of the surgical instrument and furtheraway from the surgical end effector of the surgical instrument. The term“distal” refers to the position of an element closer to the surgical endeffector of the surgical instrument and further away from the human orrobotic operator of the surgical instrument.

Also, in the following description, it is to be understood that termssuch as front, back, inside, outside, top, bottom, upper, lower and thelike are words of convenience and are not to be construed as limitingterms. Terminology used herein is not meant to be limiting insofar asdevices described herein, or portions thereof, may be attached orutilized in other orientations. The various embodiments will bedescribed in more detail with reference to the drawings.

Many surgical procedures require cutting or litigating blood vessels orother vascular tissue. With minimally invasive surgery, surgeons performsurgical operations through a small incision in the patient's body. As aresult of the limited space, surgeons often have difficulty controllingbleeding by clamping and/or tying-off transected blood vessels. Byutilizing electrosurgical instruments, such as electrosurgical forceps,a surgeon can cauterize, coagulate/desiccate, and/or simply reduce orslow bleeding by controlling the electrosurgical energy applied throughjaw members of the electrosurgical forceps, otherwise referred to asclamp arms.

Electrosurgical instruments generally comprise an electronics system forgenerating and controlling electrosurgical energy. The electronicssystem comprises an RF generation circuit to generate an RF drive signaland to provide the RF drive signal to at least one electrical contact,where the RF generation circuit also includes a resonant circuit. Theelectronics system also comprises control elements such as one or morethan one microprocessor (or micro-controller) and additional digitalelectronic elements to control the logical operation of the instrument.

The electronics elements of the power supply and RF amplifier sectionsshould be designed to have the highest efficiency possible in order tominimize heat rejected into the housing of the instrument. Efficiencyalso provides the longest storage and operational battery life possible.As described in the embodiments illustrated in FIGS. 4-16, litz wire maybe wound around a bobbin core to reduce AC losses due to high frequencyRF. The litz wire construction provides greater efficiency and thus alsoprevents heat generation in the device.

In various embodiments, efficiency of the power supply and RF drive andcontrol circuitry sections also may minimize the size of the batteryrequired to fulfill the mission life, or to extend the mission life fora given size battery. In one embodiment, the battery provides a lowsource impedance at a terminal voltage of 12.6V (unloaded) and a 1030mA-Hour capacity. Under load, the battery voltage is a nominal 11.1V,for example.

Radio frequency drive amplifier topologies may vary according to variousembodiments. In one embodiment, for example, a series resonant approachmay be employed where the operating frequency is varied to change theoutput voltage to force the medical instrument to operate according to apre-programmed load curve. In a series resonant approach, the impedanceof a series resonant network is at a minimum at the resonant frequency,because the reactance of the capacitive and inductive elements cancel,leaving a small real resistance. The voltage maximum for a seriesresonant circuit also occurs at the resonant frequency (and also dependsupon the circuit Q). Accordingly, to produce a high voltage on theoutput, the series resonant circuit should operate closer to theresonant frequency, which increases the current draw from the DC supply(e.g., battery) to feed the RF amplifier section with the requiredcurrent. Although the series resonant approach may be referred to as aresonant mode boost converter, in reality, the design is rarely operatedat the resonant frequency, because that is the point of maximum voltage.The benefit of a resonant mode topology is that if it is operated veryclose to the resonant frequency, the switching field effect transistors(FETs) can be switched “ON” or “OFF” at either a voltage or current zerocrossing, which dissipates the least amount of power in the switchingFETs as is possible.

Another feature of the RF drive and control circuitry section accordingto one embodiment, provides a relatively high turns ratio transformerwhich steps up the output voltage to about 85V RMS from the nominalbattery voltage of about 11.1V. This provides a more compactimplementation because only one transformer and one other inductor arerequired. In such a circuit, high currents are necessary on thetransformer primary to create the desired output voltage or current.Such device, however, cannot be operated at the resonant frequencybecause allowances are made to take into account for the battery voltagedropping as it is expended. Accordingly, some headroom is provided tomaintain the output voltage at the required level. A more detaileddescription of a series resonant approach is provided in commonlyassigned international PCT Patent Application No. PCT/GB2011/000778,titled “Medical Device,” filed May 20, 2011, the disclosure of which isincorporated herein by reference in its entirety.

According to another embodiment, an RF instrument topology is providedfor a handheld battery powered RF based generator for theelectrosurgical medical instrument. Accordingly, in one embodiment, thepresent disclosure provides an RF instrument topology with anarchitecture configured such that each power section of the deviceoperate at maximum efficiency regardless of the load resistancepresented by the tissue or what voltage, current, or power level iscommanded by the controller. In one embodiment, this may be implementedby employing the most efficient modalities of energy transformationpresently known and by minimizing the component size to provide a smalland light weight electronics package to fit within the instrument'shousing, for example.

In one embodiment, the RF power electronics section of the electronicssystem may be partitioned as a boost mode converter, synchronous buckconverter, and a parallel resonant amplifier. According to oneembodiment, a resonant mode boost converter section of the medicalinstrument may be employed to convert the DC battery voltage to a higherDC voltage for use by the synchronous mode buck converter. One aspect toconsider for achieving a predetermined efficiency of the resonant modeboost converter section is ratio between input and output voltages ofthe boost converter. In one embodiment, although a 10:1 ratio isachievable, the cost is that for any appreciable power on the secondarythe input currents to the boost mode transformer become quite heavy, inthe range of about 15-25A, depending on the load. In another embodimenta transformer turns ratio of about 5:1 is provided. It will beappreciated that transformer ratios in the range of about 5:1 to about10:1 also may be implemented, without limitation. In a 5:1 transformerturns ratio, the design tradeoff is managing the Q of the parallelresonant output against the boost ratio. The resonant output networkperforms two functions. First, it filters the square, digital pulsesfrom the Class D output amplifier and removes all but the fundamentalfrequency sine wave from the output. Second, it provides a passivevoltage gain due to the Q of the filter network. In other words, currentfrom the amplifier is turned into output voltage, at a gain determinedby the circuit's unloaded Q and the load resistance, which affects the Qof the circuit.

Another aspect to consider for achieving a predetermined efficiency inthe resonant mode boost converter section is to utilize a full bridgeswitcher topology, which allows half the turns ratio for the boosttransformer for the same input voltage. The tradeoff is that thisapproach may require additional FET transistors, e.g., an additional twoFETs are required over a half bridge approach, for example. Presentlyavailable switchmode FETs, however, are relatively small, and while thegate drive power is not negligible, it provides a reasonable designtradeoff.

Yet another aspect to consider for achieving a predetermined efficiencyin the resonant mode boost converter section and operating the boostconverter at maximum efficiency, is to always run the circuit at theresonant frequency so that the FETs are always switching at either avoltage or current minima, whichever is selected by the designer (ZCSvs. ZVS switching), for example. This can include monitoring theresonant frequency of the converter as the load changes, and makingadjustments to the switching frequency of the boost converter to allow35 ZVS or ZCS (Zero Voltage Switching/Zero Current Switching) to occurfor minimum power dissipation.

Yet another aspect to consider for achieving a predetermined efficiencyin the resonant mode boost converter section is to utilize a synchronousrectifier circuit instead of a conventional full-wave diode rectifierblock. Synchronous rectification employs FETs as diodes because theon-resistance of the FET is so much lower than that of even a Schottkypower diode optimized for low forward voltage drop under high currentconditions. A synchronous rectifier requires gate drive for the FETs andthe logic to control them, but offers significant power savings over atraditional full bridge rectifier.

In accordance with various embodiments, the predetermined efficiency ofa resonant mode boost converter is approximately 98-99% input to output,for example. Any suitable predetermined efficiency may be selected basedon the particular implementation. Accordingly, the embodiments describedherein are limited in this context.

According to one embodiment, a synchronous buck converter section of themedical instrument may be employed to reduce the DC voltage fed to theRF amplifier section to the predetermined level to maintain thecommanded output power, voltage or current as dictated by the loadcurve, with as little loss as is possible. The buck converter isessentially an LC lowpass filter fed by a low impedance switch, alongwith a regulation circuit to control the switch to maintain thecommanded output voltage. The operating voltage is dropped to thepredetermined level commanded by the main controller, which is runningthe control system code to force the system to follow the assigned loadcurve as a function of sensed tissue resistance. In accordance withvarious embodiments, the predetermined efficiency of a synchronous buckregulator is approximately 99%, for example. Any suitable predeterminedefficiency may be selected based on the particular implementation.Accordingly, the embodiments described herein are limited in thiscontext.

According to one embodiment, a resonant mode RF amplifier sectioncomprising a parallel resonant network on the RF amplifier sectionoutput is provided. In one embodiment, a predetermined efficiency may beachieved by a providing a parallel resonant network on the RF amplifiersection output. The RF amplifier section may be driven at the resonantfrequency of the output network, which accomplishes three things. First,the high Q network allows some passive voltage gain on the output,reducing the boost required from the boost regulator in order to producehigh voltage output levels. Second, the square pulses produced by the RFamplifier section are filtered and only the fundamental frequency isallowed to pass to the output. Third, a full-bridge amplifier isswitched at the resonant frequency of the output filter, which is to sayat either the voltage zero crossings or the current zero crossings inorder to dissipate minimum power. Accordingly, a predeterminedefficiency of the RF amplifier section is approximately 98%. Gate drivelosses may limit the efficiency to this figure or slightly lower. Anysuitable predetermined efficiency may be selected based on theparticular implementation. Accordingly, the embodiments described hereinare limited in this context.

In view of the RF instrument topology and architecture described above,an overall system efficiency of approximately 0.99*0.99*0.98, which isapproximately 96%, may be achieved. Accordingly, to deliverapproximately 45 W, approximately 1.8 W would be dissipated by theelectronics exclusive of the power required to run the main andhousekeeping microprocessors, and the support circuits such as the ADCand analog amplifiers and filters. To deliver approximately 135 W,approximately 5.4 W would be dissipated. This is the amount of powerthat would be required to implement a large jaw class generator in ahand held electrosurgical medical instrument. Overall system efficiencywould likely only be a weak function of load resistance, instead of arelatively strong one as it may be the case in some conventionalinstruments.

In various other embodiments of the electrosurgical medical instrument,a series resonant topology may be employed to achieve certainpredetermined efficiency increase by employing a full bridge amplifierfor the primary circuit and isolate the full bridge amplifier fromground to get more voltage on the primary. This provides a largerprimary inductance and lower flux density due to the larger number ofturns on the primary.

FIG. 3 illustrates an RF drive and control circuit 800, according to oneembodiment. FIG. 3 is a part schematic part block diagram illustratingthe RF drive and control circuitry 800 used in this embodiment togenerate and control the RF electrical energy supplied to theelectrosurgical instrument. As will be explained in more detail below,in this embodiment, the drive circuitry 800 is a resonant mode RFamplifier comprising a parallel resonant network on the RF amplifieroutput and the control circuitry operates to control the operatingfrequency of the drive signal so that it is maintained at the resonantfrequency of the drive circuit, which in turn controls the amount ofpower supplied to the instrument. The way that this is achieved willbecome apparent from the following description.

As shown in FIG. 3, the RF drive and control circuit 800 comprises abattery 300 arranged to supply, in this example, about 0V and about 12Vrails. An input capacitor (Cin) 802 is connected between the 0V and the12V for providing a low source impedance. A pair of FET switches 803-1and 803-2 (both of which are N-channel in this embodiment to reducepower losses) is connected in series between the 0V rail and the 30 12Vrail. FET gate drive circuitry 805 is provided that generates two drivesignals—one for driving each of the two FETs 803. The FET gate drivecircuitry 805 generates drive signals that causes the upper FET (803-1)to be on when the lower FET (803-2) is off and vice versa. This causesthe node 807 to be alternately connected to the 12V rail (when the FET803-1 is switched on) and the 0V rail (when the FET 803-2 is switchedon). FIG. 3 also shows the internal parasitic diodes 808-1 and 808-2 ofthe corresponding FETs 803, which conduct during any periods that theFETs 803 are open.

As shown in FIG. 3, the node 807 is connected to an inductor-inductorresonant circuit 810 formed by an inductor L_(s) 812 and an inductorL_(m) 814, which is the primary coil of a transformer 815. The FET gatedriving circuitry 805 is arranged to generate drive signals at a drivefrequency (f_(d)) that opens and crosses the FET switches 803 at theresonant frequency of the parallel resonant circuit 810. As a result ofthe resonant characteristic of the resonant circuit 810, the square wavevoltage at node 807 will cause a substantially sinusoidal current at thedrive frequency (f_(d)) to flow within the resonant circuit 810. Asillustrated in FIG. 9, the inductor L_(m) 814 is the primary coil of atransformer 815, the secondary coil of which is formed by inductorL_(sec) 816. The inductor L_(sec) 816 of the transformer 815 secondaryis connected to a resonant circuit 817 formed by inductor L₂, capacitorC₄ 820, capacitor C₂ 822, and capacitor C₃ 825. The transformer 815up-converts the drive voltage (V_(d)) across the inductor L_(m) 814 tothe voltage that is applied to the output parallel resonant circuit 817.The load voltage (V_(L)) is output by the parallel resonant circuit 817and is applied to the load (represented by the load resistance R_(load)819 in FIG. 3) corresponding to the impedance of the forceps' jaws andany tissue or vessel gripped by the forceps 108. As shown in FIG. 3, apair of DC blocking capacitors C_(bl1) 840-1 and C_(bl2) 840-2 isprovided to prevent any DC signal being applied to the load 819.

In one embodiment, the transformer 815 may be implemented with a CoreDiameter (mm), Wire Diameter (mm), and Gap between secondary windings inaccordance with the following specifications:

-   -   Core Diameter, D (mm)    -   D=19.9×10−3    -   Wire diameter, W (mm) for 22 AWG wire    -   W=7.366×10−4    -   Gap between secondary windings, in gap=0.125    -   G=gap/25.4

In this embodiment, the amount of electrical power supplied to theelectrosurgical instrument is controlled by varying the frequency of theswitching signals used to switch the FETs 803. This works because theresonant circuit 810 acts as a frequency dependent (loss less)attenuator. The closer the drive signal is to the resonant frequency ofthe resonant circuit 810, the less the drive signal is attenuated.Similarly, as the frequency of the drive signal is moved away from theresonant frequency of the circuit 810, the more the drive signal isattenuated and so the power supplied to the load reduces. In thisembodiment, the frequency of the switching signals generated by the FETgate drive circuitry 805 is controlled by a controller 841 based on adesired power to be delivered to the load 819 and measurements of theload voltage (V_(L)) and of the load current (I_(L)) obtained byconventional voltage sensing circuitry 843 and current sensing circuitry845. The way that the controller 841 operates will be described in moredetail below.

In one embodiment, the voltage sensing circuitry 843 and the currentsensing circuitry 845 may be implemented with high bandwidth, high speedrail-to-rail amplifiers (e.g., LMH6643 by National Semiconductor). Suchamplifiers, however, consume a relatively high current when they areoperational. Accordingly, a power save circuit may be provided to reducethe supply voltage of the amplifiers when they are not being used in thevoltage sensing circuitry 843 and the current sensing circuitry 845. Inone-embodiment, a step-down regulator (e.g., LT3502 by LinearTechnologies) may be employed by the power save circuit to reduce thesupply voltage of the rail-to-rail amplifiers and thus extend the lifeof the battery 300.

In one embodiment, the transformer 815 and/or the inductor L_(s) 812 maybe implemented with a configuration of litz wire conductors to minimizethe eddy-current effects in the windings of high-frequency inductivecomponents. These effects include skin-effect losses and proximityeffect losses. Both effects can be controlled by the use of litz wire,which are conductors made up of multiple individually insulated strandsof wire twisted or woven together. Although the term litz wire isfrequently reserved for conductors constructed according to a carefullyprescribed pattern, in accordance with the present disclosure, any wirestrands that are simply twisted or grouped together may be referred toas litz wire. Accordingly, as used herein, the term litz wire refers toany insulated twisted or grouped strands of wires.

By way of background, litz wire can reduce the severe eddy-currentlosses that otherwise limit the performance of high-frequency magneticcomponents, such as the transformer 815 and/or the inductor L_(s) 812used in the RF drive and control circuit 800 of FIG. 3. Although litzwire can be very expensive, certain design methodologies providesignificant cost reduction without significant increases in loss, ormore generally, enable the selection of a minimum loss design at anygiven cost. Losses in litz-wire transformer windings have beencalculated by many authors, but relatively little work addresses thedesign problem of how to choose the number and diameter of strands for aparticular application. Cost-constrained litz wire configurations aredescribed in C. R. Sullivan, “Cost-Constrained Selection of Strand Wireand Number in a Litz-Wire Transformer Winding,” IEEE Transactions onPower Electronics, vol. 16, no. 2, pp. 281-288, which is incorporatedherein by reference. The choice of the degree of stranding in litz wirefor a transformer winding is described in C. R. Sullivan, “OptimalChoice for Number of Strands in a Litz-Wire Transformer Winding,” IEEETransactions on Power Electronics, vol. 14, no. 2, pp. 283-291, which isincorporated herein by reference.

In one embodiment, the transformer 815 and/or the inductor L_(s) 812 maybe implemented with litz wire by HM Wire International, Inc., of Canton,Ohio or New England Wire Technologies of Lisbon, N.H., which has aslightly different construction in terms of the number of strands in theintermediate windings, but has the same total number of strands ofeither 44 gauge or 46 gauge wire by HM Wire International, Inc.Accordingly, the disclosure now turns to FIGS. 4-16, which illustrateone embodiment of the transformer 815 and the inductor L_(s) 81implemented with litz wire.

FIG. 4 illustrates a perspective view of one embodiment of thetransformer shown as transformer 815 in connection with the RF drivecircuit 800 illustrated in FIG. 3. As shown in FIG. 4, in oneembodiment, the transformer 404 comprises a bobbin 804, a ferrite core806, a primary coil 821 (e.g., inductor L_(m) 814 in FIG. 3), and asecondary coil 823 (e.g., inductor L_(sec) 816 in FIG. 3). In oneembodiment, the bobbin 804 may be a 10-pin surface mounted device (SMD)provided by Ferroxcube International Holding B.V. In one embodiment, theferrite core 806 may be an EFD 20/107 N49. In one embodiment, thetransformer 815 has a power transfer of ˜45 W, a maximum secondarycurrent of ˜1.5 A RMS, maximum secondary voltage of ˜90V RMS, maximumprimary current of ˜15.5 A RMS, and a turns ratio of 20:2 (secondaryturns:primary turns), for example. The operating frequency range of thetransformer 404 is from ˜370 kHz to ˜550 kHz, and a preferred frequencyof ˜430 kHz. It will be appreciated that these specification areprovided as examples and should not be construed to be limiting of thescope of the appended claims.

In one embodiment, the transformer 404 comprises a ferrite core materialhaving particular characteristics. The core used for both the inductor406 and the transformer 404, albeit with a different gap to yield therequired A_(L) for each component. A_(L) has units of Henrys/turns², sothe inductance of a winding may be found by using NTURNS²*A_(L). In oneembodiment, an A_(L) of 37 is used for the inductor 406, and an A_(L) of55 is used for the transformer 406. This corresponds to a gap ofapproximately 0.8 mm and 2.0 mm respectively, although the A_(L) or theinductance is the parameter to which the manufacturing process controls,with the A_(L) being an intermediate quantity that we are not measuringdirectly.

In one embodiment, the inductance of the inductor 406 and transformer404 winding may be measured directly with “golden bobbins,” which aresquarely in the middle of the tolerance bands for the windingstatistical distributions. Cores that are ground are then tested usingthe “golden bobbin” to assess whether the grind is good on the cores.Better results were yielded than the industry standard method, which isto fill a bobbin with as many windings as they can fit on the bobbin,and then back calculating the A_(L) of the core, and controlling A_(L)instead of the inductance. It was found that using a “golden bobbin” inthe manufacturing process yielded better results. The bobbin is what thecopper windings are secured to, and the ferrite E cores slip through ahole in the bobbin, and are secured with clips.

FIG. 5 illustrates a perspective view of one embodiment of the primarycoil 821 (e.g., inductor L_(m) 814 in FIG. 3) of the transformer 404illustrated in FIG. 4. In one embodiment, the primary coil 821 windingsmay be constructed using 300 strand/46 gauge litz wire as indicated inTABLE 1 below, among other suitable configurations. In one embodiment,primary coil 821 has an inductance of ˜270 nH, an AC resistance <46mΩ,and a DC resistance of ≤5 mΩ, for example.

TABLE 1 Primary Coil 821 (Lm 814) 46 Gauge Litz Wire 300 Strands 46 AWG-24 turns per foot (TPF) Single Build MW80 155° C. Single Nylon ServedConstruction: 5 × 3 × 20/46 AWG Ft per lb: 412 Nominal OD: 0.039″Nominal

FIG. 7 illustrates a bottom view of the primary coil 821 (e.g., inductorL_(m) 814 in FIG. 3) illustrated in FIG. 5. FIG. 8 illustrates a sideview of the primary coil 821 illustrated in FIG. 5. FIG. 9 illustrates asectional view of the primary coil 821 illustrated in FIG. 5 taken alongsection 28-28.

FIG. 6 illustrates a perspective view of one embodiment of a secondarycoil 823 (e.g., inductor L_(sec) 816 in FIG. 3) of the transformer 404illustrated in FIG. 4. In one embodiment, the secondary coil 823windings may be constructed using 105 strand/44 gauge litz wire asindicated in TABLE 2 below, among other suitable configurations. In oneembodiment, the secondary coil 823 has an inductance of 22 μH±5% @430kHz, an AC resistance <2.5Ω, and a DC resistance ≤80 mΩ, for example.

TABLE 2 Secondary Coil 823 (Lsec 816) 44 Gauge Litz Wire 105 Strands 44AWG 24 TPF Single Build MW80 155° C. Single Nylon Served Construction: 5× 21/44 AWG Ft per lb: 1214 Nominal OD: 0.023″ Nominal

FIG. 10 illustrates a bottom view of the secondary coil 823 (e.g.,inductor L_(sec) 816 in FIG. 3) illustrated in FIG. 6. FIG. 11illustrates a side view of the secondary coil 823 illustrated in FIG. 6.FIG. 12 illustrates a sectional view of the secondary coil 8235illustrated in FIG. 11 taken along section 31-31.

FIG. 13 is a perspective view of one embodiment of the inductor 406shown as inductor L_(s) 812 in connection with the RF drive circuit 800illustrated in FIG. 3. As shown in FIG. 13, in one embodiment, theinductor 406 comprises a bobbin 809, a ferrite core 811, and a coil 813.In one embodiment, the bobbin 809 may be a 10-pin surface mounted device(SMD) provided by Ferroxcube International Holding B.V. In oneembodiment, the ferrite core 811 may be an EFD 20/107 N49. In oneembodiment, the coil 813 windings may be constructed using 300 strand/46gauge litz wire wound at 24 TPF. In one embodiment, the inductor L_(s)812 may have an inductance of ˜345 nH±6% @430 kHz, an AC resistance <50mΩ, and a DC resistance ≤7 mΩ, for example. The operating frequencyrange of the inductor L_(s) 812 is from ˜370 kHz to ˜550 kHz, and apreferred frequency of ˜430 kHz, and an operating current of ˜15.5 ARMS. It will be appreciated that these specification are provided asexamples and should not be construed to be limiting of the scope of theappended claims.

FIG. 14 illustrates a bottom view of the inductor 406 (e.g., inductorL_(s) 812 in FIG. 3) illustrated in FIG. 13. FIG. 15 illustrates a sideview of the inductor 406 illustrated in FIG. 13. FIG. 16 illustrates asectional view of the inductor 406 illustrated in FIG. 15 taken alongsection 35-35.

Accordingly, as described above in connection with FIGS. 4-16, in oneembodiment, the transformer 404 (e.g., transformer 815) and/or theinductor 406 (e.g., inductor 812) used in the RF drive and controlcircuit 800 of FIG. 3 may be implemented with litz wire. One litz wireconfiguration may be produced by twisting 21 strands of 44 AWG SPN wireat 18 twists per foot (left direction twisting). Another litz wireconfiguration may be produced by twisting 5×21/44 AWG (105/44 AWG SPN),also at 18 twists per foot (left direction twisting). Other litz wireconfigurations include 300/46 AWG litz wire as well as 46 AWG or finergauge size wire.

FIG. 17 illustrates the main components of the controller 841, accordingto one embodiment. In the embodiment illustrated in FIG. 17, thecontroller 841 is a microprocessor based controller and so most of thecomponents illustrated in FIG. 3 are software based components.Nevertheless, a hardware based controller 841 may be used instead. Asshown, the controller 841 includes synchronous I, Q sampling circuitry851 that receives the sensed voltage and current signals from thesensing circuitry 843 and 845 and obtains corresponding samples whichare passed to a power, V_(rms) and I_(rms) calculation module 853. Thecalculation module 853 uses the received samples to calculate the RMSvoltage and RMS current applied to the load 819 and, from the voltageand current, the power that is presently being supplied to the load 839.The determined values are then passed to a frequency control module 855and a medical device control module 857. The medical device controlmodule 857 uses the values to determine the present impedance of theload 819 and based on this determined impedance and a pre-definedalgorithm, determines what set point power (P_(set)) should be appliedto the frequency control module 855. The medical device control module857 is in turn controlled by signals received from a user input module859 that receives inputs from the user and also controls output devices(lights, a display, speaker or the like) on the handle of the instrumentvia a user output module 861.

The frequency control module 855 uses the values obtained from thecalculation module 853 and the power set point (P_(set)) obtained fromthe medical device control module 857 and predefined system limits (tobe explained below), to determine whether or not to increase or decreasethe applied frequency. The result of this decision is then passed to asquare wave generation module 863 which, in this embodiment, incrementsor decrements the frequency of a square wave signal that it generates by1 kHz, depending on the received decision. As those skilled in the artwill appreciate, in an alternative embodiment, the frequency controlmodule 855 may determine not only whether to increase or decrease thefrequency, but also the amount of frequency change required. In thiscase, the square wave generation module 863 would generate thecorresponding square wave signal with the desired frequency shift. Inthis embodiment, the square wave signal generated by the square wavegeneration module 863 is output to the FET gate drive circuitry 805,which amplifies the signal and then applies it to the FET 803-1. The FETgate drive circuitry 805 also inverts the signal applied to the FET803-1 and applies the inverted signal to the FET 803-2.

FIG. 18 is a signal plot illustrating the switching signals applied tothe FETs 803, a sinusoidal signal representing the measured current orvoltage applied to the load 819, and the timings when the synchronoussampling circuitry 851 samples the sensed load voltage and load current,according to one embodiment. In particular, FIG. 18 shows the switchingsignal (labeled PWM1 H) applied to upper FET 803-1 and the switchingsignal (labeled PWM1 L) applied to lower FET 803-2. Although notillustrated for simplicity, there is a dead time between PWM1 H and PWM1L to ensure that that both FETs 803 are not on at the same time. FIG. 18also shows the measured load voltage/current (labeled OUTPUT). Both theload voltage and the load current will be a sinusoidal waveform,although they may be out of phase, depending on the impedance of theload 819. As shown, the load current and load voltage are at the samedrive frequency (f_(e)) as the switching Signals (PWM1 H and PWM1 L)used to switch the FETs 803. Normally, when sampling a sinusoidalsignal, it is necessary to sample the signal at a rate corresponding toat least twice the frequency of the signal being sampled—i.e. twosamples per period. However, as the controller 841 knows the frequencyof the switching signals, the synchronous sampling circuit 851 cansample the measured voltage/current signal at a lower rate. In thisembodiment, the synchronous sampling circuit 851 samples the measuredsignal once per period, but at different phases in adjacent periods. InFIG. 18, this is illustrated by the “I” sample and the “Q” sample. Thetiming that the synchronous sampling circuit 851 makes these samples iscontrolled, in this embodiment, by the two control signals PWM2 andPWM3, which have a fixed phase relative to the switching signals (PWM1 Hand PWM1 L) and are out of phase with each other (preferably by quarterof the period as this makes the subsequent calculations easier). Asshown, the synchronous sampling circuit 851 obtains an “I sample onevery other rising edge of the PWM2 signal and the synchronous samplingcircuit 851 obtains a “0” sample on every other rising edge of the PWM3signal. The synchronous sampling circuit 851 generates the PWM2 and PWM3control signals from the square wave signal output by the square wavegenerator 863 (which is at the same frequency as the switching signalsPWM1 H and PWM1 L). Thus control signals PWM2 and PWM3 also change(whilst their relative phases stay the same). In this way, the samplingcircuitry 851 continuously changes the timing at which it samples thesensed voltage and current signals as the frequency of the drive signalis changed so that the samples are always taken at the same time pointswithin the period of the drive signal. Therefore, the sampling circuit851 is performing a “synchronous” sampling operation instead of a moreconventional sampling operation that just samples the input signal at afixed sampling rate defined by a fixed sampling clock.

The samples obtained by the synchronous sampling circuitry 851 are thenpassed to the power, V_(rms) and I_(rms) calculation module 853 whichcan determine the magnitude and phase of the measured signal from justone “I” sample and one “Q” sample of the load current and load voltage.However, in this embodiment, to achieve some averaging, the calculationmodule 853 averages consecutive “I” samples to provide an average “I”value and consecutive “Q” samples to provide an average “0” value; andthen uses the average I and Q values to determine the magnitude andphase of the measured signal (in a conventional manner). As thoseskilled in the art will appreciate, with a drive frequency of about 400kHz and sampling once per period means that the synchronous samplingcircuit 851 will have a sampling rate of 400 kHz and the calculationmodule 853 will produce a voltage measure and a current measure every0.01 ms. The operation of the synchronous sampling circuit 851 offers animprovement over existing products, where measurements can not be madeat the same rate and where only magnitude information is available (thephase information being lost).

In one embodiment, the RF amplifier and drive circuitry for theelectrosurgical medical instrument employs a resonant mode step-upswitching regulator, running at the desired RF electrosurgical frequencyto produce the required tissue effect. The waveform illustrated in FIG.18 can be employed to boost system efficiency and to relax thetolerances required on several custom components in the electronicssystem 400. In one embodiment, a first generator control algorithm maybe employed by a resonant mode switching topology to produce the highfrequency, high voltage output signal necessary for the medicalinstrument. The first generator control algorithm shifts the operatingfrequency of the resonant mode converter to be nearer or farther fromthe resonance point in order to control the voltage on the output of thedevice, which in turn controls the current and power on the output ofthe device. The drive waveform to the resonant mode converter hasheretofore been a constant, fixed duty cycle, with frequency (and notamplitude) of the drive waveform being the only means of control.

FIG. 19 illustrates a drive waveform for driving the FET gate drivecircuitry 805, according to one embodiment. Accordingly, in anotherembodiment, a second generator control algorithm may be employed by aresonant mode switching topology to produce the high frequency, highvoltage output signal necessary for the medical instrument. The secondgenerator control algorithm provides an additional means of control overthe amplifier in order to reduce power output in order for the controlsystem to track the power curve while maintaining the operationalefficiency of the converter. As shown in FIG. 19, according to oneembodiment, the second generator control algorithm is configured to notonly modulate the drive frequency that the converter is operating at,but to also control the duty cycle of the drive waveform by duty cyclemodulation. Accordingly, the drive waveform 890 illustrated in FIG. 19exhibits two degrees of freedom. Advantages of utilizing the drivewaveform 890 modulation include flexibility, improved overall systemefficiency, and reduced power dissipation and temperature rise in theamplifier's electronics and passive inductive components, as well asincreased battery life due to increased system efficiency.

RF Amplifier Topology

FIG. 1 illustrates one embodiment of an RF amplifier 100 with one ormore of taps on the primary coil 104, wherein each tap is controlled bya half bridge driver 108. As discussed below, each half bridge driver108 may comprise transistors, MOSFETs, insulated-gate bipolartransistors (IGBTs) or any other suitable switching devices configuredin a half bridge drive configuration. The output transformer primarywinding 104 may be driven between any two of the half bridge drivers108, with the number of turns between the half bridge drivers 108 andthe fixed output winding 106 determining the overall turns ratio for thetransformer 102.

This topology allows for a lower turns ration for high current output onthe secondary coil 106 while limiting the primary current to a valuethat is compatible with currently available lithium-ion (Li-Ion)batteries. For example, a primary current in the range of 20-30 Aimplies a turns ration of about 4:1. Conversely, when generating arelatively high voltage on the secondary coil 106, for example in therange of 170-250V RMS, it is desirable to have a relatively higher turnsration between the primary coil 104 and the secondary coil 106, forexample a ration of about 15:1. This can be accomplished by reducing thenumber of turns in the primary coil 104 relative to a secondary coil 106with a fixed number of turns.

This topology provides the ability to dynamically vary the turns ratio,in real time and in sync the output waveform being generated. This is inkeeping with a zero-voltage switching (ZVS) or zero current switching(ZCS) methodology for driving the amplifier at its resonant frequencyfor maximum efficiency and minimum power dissipation in the switchingtransistors, MOSFETs, IGBTs or other switching devices.

With this topology, the amplifier 100 can also be dynamically driven ina half bridge or a full bridge mode on a output-cycle-by-cycle basis,within a resonant mode drive scheme. This allows for better outputregulation.

This topology also provides the ability to match the turns ratio to theregion of tissue resistance. This optimizes the losses in thetransformer windings by preventing excessive currents in the primarycoil 104. It also optimizes the battery voltage required to produce ahigh voltage on the secondary coil 106 for large-jaw devices and thetypes of anatomical structures such devices are typically called upon toseal and cut. Multiple turns ratio values may be provided in order tooptimize each region or sub-region of the electrosurgical power curve,as necessary.

With this topology the efficiency of the amplifier may be keptarbitrarily close to the optimal value by selection of taps on theprimary coil 104.

An arbitrary number of half bridge driver 108 circuits and transformertaps may be provided, tailored to the performance requirements of theparticular RF amplifier 100.

FIG. 2 illustrates one embodiment of a half bridge circuit 110 that maybe employed by the half bridge driver 108. The half bridge circuit 110comprises an upper switch 112 and a lower switch 114, here illustratedas MOSFETs, wherein the upper 112 and lower 114 switches are connectedin a cascade arrangement. The half bridge circuit 110 further comprisesan input voltage +VBatt that is provided by the instrument's onboardbatteries and a ground return. The half bridge circuit 110 furthercomprises a high-side gate drive input 116 and a low-side gate driveinput 118. The output from the half bridge circuit 110 is at the nodebetween the upper 112 and lower 114 switches. In operation, the switches112, 114 are turned on and off complementary to each other, withnon-overlapping dead time, by applying the correct voltage waveforms ateach of the gate drive inputs. This half bridge circuit 110 topologyprovides for four-quadrant switching, zero-voltage switching (ZVS),zero-current switching (ZCS), high-frequency operation, low EMI, andhigh efficiency.

While FIG. 2 illustrates a half bridge circuit 110 comprising MOSFETswitches, any switch may be used, such as for example transistors, IGBTsor any other suitable switching device.

Furthermore, the design of half bridge circuits is well understood, andany half bridge circuit may be employed in the amplifier topologydescribed above.

Various embodiments of the amplifier 100 as described above may comprisealternate topologies. For example, some embodiments may use solid-stateswitching elements, such as MOSFETs or other semiconductors that can besimilarly controlled. Other embodiments may use physical relays, thoughphysical relays have limitations, including relatively long switchingtimes and arcing that occurs at the contacts when they are switched,caused because it is not possible to switch mechanical relays at acurrent or voltage zero crossing at the primary coil 104 or secondarycoil 106. Arcing is an issue for designs that are intended to bereprocessed and reused.

FIG. 1 illustrates on embodiment of an amplifier 100 with a parallelresonant output section, comprising a parallel capacitor 120. Such anamplifier 100 is intended to produce a relatively pure sine wave as anoutput. The parallel capacitor 120 that forms the other half of theresonant tank circuit, with the transformer secondary inductance andleakage inductance, may be omitted, to produce a square wave approximateoutput waveform. when a peak-detector type of circuit is used for outputvoltage and current sensing, then issues with higher Nyquis ratesampling on the output may be avoided: the output pseudo-square wavewill contain significant energy at the third, fifth, seventh and ninthharmonics, which will distort the output measurements if the A/Dconverter in the design does not have adequate bandwidth to sample theseharmonics and keep them from aliasing. A lowpass anti-aliasing filterand software linearization (correction) for measured quantities couldalso be contemplated as solutions to reduce the Nyquist rate samplingrate required to avoid aliasing.

It is worthy to note that any reference to “one aspect,” “an aspect,”“one embodiment,” or “an embodiment” means that a particular feature,structure, or characteristic described in connection with the aspect isincluded in at least one aspect. Thus, appearances of the phrases “inone aspect,” “in an aspect,” “in one embodiment,” or “in an embodiment”in various places throughout the specification are not necessarily allreferring to the same aspect. Furthermore, the particular features,structures or characteristics may be combined in any suitable manner inone or more aspects.

Although various embodiments have been described herein, manymodifications, variations, substitutions, changes, and equivalents tothose embodiments may be implemented and will occur to those skilled inthe art. Also, where materials are disclosed for certain components,other materials may be used. It is therefore to be understood that theforegoing description and the appended claims are intended to cover allsuch modifications and variations as falling within the scope of thedisclosed embodiments. The following claims are intended to cover allsuch modification and variations.

Although various embodiments have been described herein, manymodifications, variations, substitutions, changes, and equivalents tothose embodiments may be implemented and will occur to those skilled inthe art. Also, where materials are disclosed for certain components,other materials may be used. It is therefore to be understood that theforegoing description and the appended claims are intended to cover allsuch modifications and variations as falling within the scope of thedisclosed embodiments. The following claims are intended to cover allsuch modification and variations.

What is claimed is:
 1. A control circuit for a radio frequency drive ofan electrosurgical device, the control circuit comprising: a voltagedata input configured to receive voltage data from a voltage sensingcircuit; a current data input configured to receive current data from acurrent sensing circuit; and a switching signal output configured tosource a switching signal to the radio frequency drive of theelectrosurgical device, wherein the control circuit is configured toadjust a frequency of the switching signal based on the voltage data andthe current data, and wherein the radio frequency drive comprising atransformer, wherein the transformer comprises: a first tap including afirst half bridge driver; a second tap including a second half bridgedriver; a third tap including a third half bridge driver; a firstportion of a primary coil located between the first tap and the secondtap; a second portion of the primary coil located between the second tapand the third tap; and a secondary coil, wherein the first, second, andthird half bridge drivers are configured to selectively turn on or turnoff the first, second, and third taps, respectively, wherein two of thefirst, second, and third taps are selected to drive the primary coilbetween the two selected taps, which allows the transformer to provide aplurality of winding ratio values, wherein a number of coil turns of theprimary coil between the two selected taps and a number of coil turns ofthe secondary coil determine an overall winding ratio value of thetransformer, wherein the overall winding ratio value is one of theplurality of winding ratio values provided by the transformer.
 2. Thecontrol circuit of claim 1, further comprising a synchronous I/Qsampling circuit, configured to sample the voltage data and to samplethe current data.
 3. The control circuit of claim 2, wherein thesynchronous I/Q sampling circuit is configured to sample the voltagedata and the current data based on a period of the switching signal. 4.The control circuit of claim 3, wherein the synchronous I/Q samplingcircuit is configured to sample the voltage data and to sample thecurrent data once per period of the switching signal at a first phase ofa first period of the switching signal, and at a second phase of anadjacent period of the switching signal.
 5. The control circuit of claim3, wherein the synchronous I/Q sampling circuit is configured to samplethe voltage data and to sample the current data in-phase with andin-quadrature to the switching signal.
 6. The control circuit of claim5, further comprising a power calculation circuit configured tocalculate a power value, an RMS voltage value, and an RMS current valuebased on the sampled voltage data and the sampled current data.
 7. Thecontrol circuit of claim 6, wherein the power calculation circuit isconfigured to calculate the RMS voltage value from an average ofsuccessive in-phase samples of the voltage data and an average ofsuccessive in-quadrature samples of the voltage data, and wherein thepower calculation circuit is configured to calculate the RMS currentvalue from an average of successive in-phase samples of the current dataand an average of successive in-quadrature samples of the current data.8. The control circuit of claim 6, further comprising a square wavegeneration module in data communication with the switching signal outputand is configured to generate the switching signal.
 9. The controlcircuit of claim 8, wherein the square wave generation module has afrequency resolution of 1 kHz.
 10. The control circuit of claim 8,wherein the square wave generation module is configured to adjust afrequency of the switching signal based at least in part on the RMSvoltage value, the RMS current value, and the power value.
 11. Thecontrol circuit of claim 10, wherein the square wave generation moduleis configured to increase a frequency of the switching signal ordecrease the frequency of the switching signal.
 12. A method ofcontrolling a radio frequency drive circuit of an electrosurgicaldevice, the method comprising: sampling, by a synchronous I/Q samplingcircuit, a plurality of output voltage values of an output voltage ofthe electrosurgical device; sampling, by the synchronous I/Q samplingcircuit, a plurality of output current values of an output current ofthe electrosurgical device; and sourcing, by a square wave generationmodule, a switching signal to drive the radio frequency drive circuit,wherein the output voltage and the output current are sourced by theradio frequency drive circuit comprising a transformer, wherein thetransformer further comprises: a first tap including a first half bridgedriver; a second tap including a second half bridge driver; a third tapincluding a third half bridge driver; a first portion of a primary coillocated between the first tap and the second tap; a second portion ofthe primary coil located between the second tap and the third tap; and asecondary coil, wherein the first, second, and third half bridge driversare configured to selectively turn on or turn off the first, second, andthird taps, respectively, wherein two of the first, second, and thirdtaps are selected to drive the primary coil between the two selectedtaps, which allows the transformer to provide a plurality of windingratio values, wherein a number of coil turns of the primary coil betweenthe two selected taps and a number of coil turns of the secondary coildetermine an overall winding ratio value of the transformer, wherein theoverall winding ratio value is one of the plurality of winding ratiovalues provided by the transformer.
 13. The method of claim 12 furthercomprising, adjusting, by the square wave generation module, a frequencyof the switching signal.
 14. The method of claim 13, wherein adjusting,by the square wave generation module, a frequency of the switchingsignal comprises adjusting, by the square wave generation module, afrequency of the switching signal to a resolution of 1 kHz.
 15. Themethod of claim 13, wherein adjusting, by the square wave generationmodule, a frequency of the switching signal comprises adjusting, by thesquare wave generation module, a frequency of the switching signal basedat least in part on the plurality of sampled output voltage values andthe plurality of sampled output current values.
 16. The method of claim12, wherein sampling, by a synchronous I/Q sampling circuit, a pluralityof output voltage values comprises sampling, by the synchronous I/Qsampling circuit, a plurality of output voltage values based on a periodof the switching signal, and wherein sampling, by a synchronous I/Qsampling circuit, a plurality of output current values comprisessampling, by the synchronous I/Q sampling circuit, a plurality of outputcurrent values based on the period of the switching signal.
 17. Themethod of claim 12, wherein sampling, by a synchronous I/Q samplingcircuit, a plurality of output voltage values comprises sampling, by thesynchronous I/Q sampling circuit, a plurality of output voltage valuesin-phase with and in-quadrature to the switching signal, and whereinsampling, by a synchronous I/Q sampling circuit, a plurality of outputcurrent values comprises sampling, by a synchronous I/Q samplingcircuit, a plurality of output current values in-phase with andin-quadrature to the switching signal.
 18. The method of claim 12,further comprising calculating, by a power calculation circuit, an RMSvoltage value, and an RMS current value based on the output voltagevalues and the output current values.
 19. The method of claim 18,further comprising, adjusting, by the square wave generation module, afrequency of the switching signal based at least in part on the RMSvoltage value and the RMS current value.